Apparatus and method for fixed-frequency control in a switching power supply

ABSTRACT

A sliding-mode switching power supply ( 24 ) having N phases ( 28 ) and a method of operating the power supply ( 24 ) are provided. N switches ( 30 ) are coupled to a bipolar power source ( 22 ), with each switch ( 30 ) effecting one phase ( 28 ). An inductance ( 32 ) is coupled to each switch ( 30 ), and a capacitance ( 36 ) is coupled to the inductances ( 32 ). A load ( 26 ) is coupled across the capacitance ( 36 ). A monitor circuit ( 38 ) is coupled to the inductances ( 32 ) and the capacitance ( 36 ) and configured to monitor currents (I L ) through the inductances ( 32 ) and/or a voltage (V C ) across the capacitance ( 36 ). A sliding-surface generator ( 78 ) is coupled to the monitor circuit ( 38 ) and generates a single sliding surface (σ) for all phases ( 28 ). A constant-frequency control ( 104 ) forms a variable window (Δσ) for the sliding surface (σ). A switching circuit ( 138 ) switches the switches ( 30 ) at a switching frequency (f S ) determined by the variable window (Δσ). The constant-frequency control ( 104 ) adjusts the variable window (Δσ) to maintain the switching frequency (f S ) substantially constant.

RELATED INVENTIONS

The present invention is a continuation of “Apparatus And Method ForFixed-Frequency Control In A Switching Power Supply,” Ser. No.10/962,823, filed 7 Oct. 2004, now U.S. Pat. No. 7,091,708 which claimsbenefit under 35 U.S.C. 119(e) to “Switching Power Supply withSliding-Mode Control,” U.S. Provisional Patent Application Ser. No.60/588,098, filed 15 Jul. 2004, which is incorporated by referenceherein.

The present invention is related to the following U.S. patentapplications, each of which was filed on the same date as theapplication from which this application is a continuation, is assignedto the assignee hereof, and is incorporated by reference herein.

“Apparatus and Method for Sliding-Mode Control in a Multiphase SwitchingPower Supply,” by Zaki Moussaoui, Brian L. Allen, and Larry G. Pearce,U.S. patent application Ser. No. 10/961,950;

“Apparatus and Method for State-Variable Synthesis in a Switching PowerSupply,” by Zaki Moussaoui, U.S. patent application Ser. No. 10/961,439;and

“Apparatus and Method for Transient Control in a Multiphase SwitchingPower Supply,” by Zaki Moussaoui and Thomas Victorin, U.S. patentapplication Ser. No. 10/962,088.

TECHNICAL FIELD OF THE INVENTION

The present invention relates to the field of switching power supplies.More specifically, the present invention relates to the field ofswitching power supplies that utilize a comprehensive feedback signaland fixed-frequency control.

BACKGROUND OF THE INVENTION

Modern electronic equipment often requires low-ripple, high-currentpower sources at low to moderate voltages. Conventional switching powersupplies can meet these requirements. In addition, switching powersupplies are typically more efficient, lighter, and less expensive thantheir traditional analog counterparts, all of which are advantages inthe modern world.

FIG. 1 shows a simplified schematic diagram of a single-phase system 10utilizing a conventional buck-converter type of switching power supply11. Power supply 11 incorporates a double-throw switch 12. Switch 12couples to an inductance 13, and alternately connects a first node ofinductance 13 to an input D-C power source 14 and a ground (common) eachtime switch 12 is toggled. A capacitance 15 and a load 16 are coupled inparallel between a second node of inductance 13 and ground.

Switch 12 is typically realized as a pair of MOSFETs or other activedevices operating as double-throw switch 12, and makes a connection ineither throw. For the sake of convention, however, this discussion willassume that switch 12 is “on” when it connects inductance 13 to powersource 14 and “off” when it connects inductance 13 to ground.

When switch 12 is on, current flows into inductance 13. The energycontained in inductance 13 increases. Current flows from inductance 13into capacitance 15 and load 16. The energy contained in capacitance 15also increases. Load 16 receives its energy primarily from inductance13.

When switch 12 is off, current flows from inductance 13 to ground. Theenergy contained in inductance 13 decreases. Current flows fromcapacitance 15 into load 16. Load 16 receives its energy primarily fromcapacitance 15.

A monitor circuit 17 monitors state variables, such as a voltage acrosscapacitance 15 and a current through inductance 13, to determine when totoggle switch 12. A control circuit 18 controls the switching of switch12 in response to the state variables monitored by monitor circuit 17.

FIG. 2 shows a simplified schematic diagram of an extension ofsingle-phase system 10 into a multiphase system 10. The followingdiscussion refers to FIG. 2 except as noted therein.

In FIG. 2, system 10 utilizes a multiphase version of switching powersupply 11. Often, the high current and/or low ripple requirements ofload 16 may be such as to exceed the capabilities of a single switch 12.In such situations, it is common to use N switches 12, where N is aninteger greater than one. Each of the N switches 12 couples to its ownone of N inductances 13, and connects that one inductance 13 to powersource 14 when that switch 12 is on, and to ground when that switch 12is off. A single power source 14 may be common to all N switches 12.Normally, all inductances 13 couple to a single capacitance 15 and load16.

Multiphase power supply 11 has N phases 19, where each switch 12 effectsone of the N phases 19. Each of the N phases 19 is interleaved with theothers. The power dissipated by each switch 12 is a function of the dutycycle of that switch 12. The duty cycle of a given switch 12 istypically maintained at no more than 1/N with N interleaved phases 19.Putting it another way, a symmetrical multiphase system 10 wouldtypically provide approximately N times the current of a single-phasesystem using the same components for switches 12.

In concept, therefore, there is a significant advantage to multiphasesystem 10 with a large number of phases 19. However, problems existswith such systems 10 in that, as the number of phases 19 increases,control circuit 18 increases in complexity in order to control andmaintain the timing of phases 19. This increase in complexity isreflected in a decrease in reliability and an increase in cost.

One such problem is that each of the N phases 19 should ideally provideapproximately the same current. The use of components having typicaltolerances may nevertheless result in a wide difference in currentsbetween phases 19, and may result in one switch 12 carrying excessivecurrent. This necessitates that a typical control circuit 18 must managethe individual phase currents, as well as the collective current and thephase timing.

Conventionally, a linear controller is used for control circuit 18. Thisis a complex circuit requiring inputs from at least N+1 state variables.Moreover, the parameters of a linear controller are tightly matched withthe parameters of inductances 13, capacitance 15, and load 16. Thisoften necessitates a change in the controller itself whenever there iseven a slight change in inductances 13, capacitance 15, and/or load 16.Consequently, costs associated with control circuit 18 when realized asa linear controller may initially be undesirably high and may beexacerbated by the inability of control circuit 18 to accommodatechanges in inductances 13, capacitance 15, and/or load 16.

Control circuit 18 may be realized as a hysteretic controller.Conventional implementations of hysteretic controllers, however, areunsuitable for multiphase systems 10. Even in single-phase systems,hysteretic controllers characteristically exhibit poor performance. Thispoor performance is due, at least in part, to the inherent lag betweenthe voltage across capacitance 15 and the current through inductances13. In addition, the switching frequency of switch 12 is dependent uponload 16. That is, the switching frequency will vary as load 16 varies.

Control circuit 18 for system 10 may also be realized as a sliding-modecontroller, which may also be viewed as a form of second-orderhysteretic controller. Conventional implementations of sliding-modecontrollers are also considered unsuitable for multiphase systems 10,but might offer improvements in performance over hysteretic controllersin single-phase systems. With conventional sliding-mode controllers,however, the switching frequency is still dependent upon load 16.

Moreover, simply scaling hysteretic or sliding-mode controllers tomanage the phase currents of inductances 13, the collective current, andthe phase timing for switches 12 in multiphase system 10 produces nosignificant improvement in complexity over liner controllers, and doesnot address the problems of reliability and cost.

Conventional hysteretic and sliding-mode control circuits 18 used inconventional power supplies 11 have switching frequencies that are afunction of load 16. That means, as load 16 changes, the switchingfrequency changes. Since a ripple frequency across capacitance 15, andhence across load 16, is directly related to the switching frequency,changes in load 16 bring about changes in the ripple frequency. Theripple frequency present at load 16 may cause harmonic and/orintermodulation interference with whatever electronic device serves asload 16. Were the ripple frequency to be constant, then the ripplefrequency may be chosen to exist in an area of the spectrum to whichload 16 is insensitive. Alternatively, relatively simple filtrationwithin load 16 may be used to suppress the effects of the ripplefrequency. Allowing ripple frequency to vary makes it difficult toignore or suppress these effects.

Another problem exists with conventional multiphase switching powersupplies utilizing either a hysteretic or sliding-mode control circuit18 in that, under certain conditions, sudden shifts in load 16 may causea given switch 12 to enter a lockup condition, i.e., to remain on for anexcessive length of time. Under such circumstances, that switch 12 is indanger of exceeding its tolerances and failing. Specifically, thecurrent through that switch 12 for that excessive length of time maycause that switch 12 to exceed its power rating, and may thereby cause acatastrophic failure of that switch 12.

There is a need, therefore, for a switching power supply that has acontrol circuit that is simple, reliable, and inexpensive, requires aminimal number of state variables, maintains substantially equal currentthrough all inductances, is substantially independent of the tolerancesof its components, is immune to variations in the load, is tolerant ofswitch lockup conditions, and is suitable for either single-phase ormultiphase systems.

SUMMARY OF THE INVENTION

Accordingly, it is an advantage of the present invention that anapparatus and method for fixed-frequency control in a switching powersupply are provided.

It is another advantage of the present invention that a switching powersupply is provided that has a control circuit that is simple, reliable,and inexpensive.

It is another advantage of the present invention that a switching powersupply is provided that requires no more than two state variables,regardless of the number of phases.

It is another advantage of the present invention that a switching powersupply is provided that is substantially independent of componenttolerance.

It is another advantage of the present invention that a power supply isprovided that incorporates a control circuit having a switchingfrequency relatively immune to variations in the load.

It is another advantage of the present invention that a switching powersupply is provided that is suitable for either single-phase ormultiphase systems.

The above and other advantages of the present invention are carried outin one form by a method of operating a fixed-frequency switching powersupply N phases, where N is a positive integer. The method incorporatesgenerating a comprehensive feedback signal from no more than two statevariables of the power supply, forming a variable window for thecomprehensive feedback signal, translating the comprehensive feedbacksignal into a stream of switching pulses, switching N switches inresponse to the stream of switching pulses, and effecting one of the Nphases with each of the N switches.

The above and other advantages of the present invention are carried outin another form by a fixed-frequency switching power supply having Nphases, where N is a positive integer. The power supply includes Nswitches configured to be coupled to a bipolar power source, Ninductances, wherein each of the N inductances is coupled to one of theN switches, a capacitance coupled to each of the N inductances andconfigured to be coupled across a load, a feedback-signal generatorcoupled to the capacitance and configured to generate a comprehensivefeedback signal in response to from no more than two state variables ofthe power supply, a pulse-width-modulation (PWM) generator coupled tothe comprehensive feedback-signal generator and configured to translatethe feedback signal into a stream of switching pulses at a substantiallyconstant switching frequency, and a phase selector coupled to the Nswitches, coupled to the PWM generator, and configured to switch the Nswitches at the substantially constant switching frequency in responseto the stream of switching pulses so that each of the N switches effectsone of the N phases.

BRIEF DESCRIPTION OF THE DRAWINGS

A more complete understanding of the present invention may be derived byreferring to the detailed description and claims when considered inconnection with the Figures, wherein like reference numbers refer tosimilar items throughout the Figures, and.

FIG. 1 shows a simplified schematic diagram of a single-phase systemutilizing a prior-art switching power supply;

FIG. 2 shows a simplified schematic diagram of a an extension of theprior-art system of FIG. 1 into a multiphase system;

FIG. 3 shows a simplified schematic diagram of a single-phase ormultiphase system in accordance with a preferred embodiment of thepresent invention;

FIG. 4 shows a block diagram of a monitor circuit for the power supplyof FIG. 3 in accordance with a preferred embodiment of the presentinvention;

FIG. 5 shows a block diagram of a control circuit for the power supplyof FIG. 3 in accordance with a preferred embodiment of the presentinvention;

FIG. 6 shows a phase plot depicting a sliding surface and a variablewindow therefor in accordance with a preferred embodiment of the presentinvention; and

FIG. 7 shows a chart depicting the switching of each of N switches forthe N phases of the power supply of FIG. 3 in accordance with apreferred embodiment of the present invention.

DETAILED DESCRIPTION OF THE PREFERRED EMBODIMENTS

FIG. 3 shows a simplified schematic diagram of a multiphase system 20including a D-C power source 22 coupled to a switching power supply 24,which is coupled to a load 26. The following discussion refers to FIG.3.

System 20 is made up of power source 22, power supply 24, and load 26.Power source 22 is configured to provide D-C energy in a first form asinput energy to power supply 24. This input energy consists of an inputvoltage V_(In) at an input current I_(In). Power source 22 may be abattery, an A-C to D-C converter, a solar array, a generator, analternator, or any other source of suitable D-C energy.

Load 26 demands D-C energy in a second form as output energy from powersupply 24. This output energy consists of an output voltage V_(Out) atan output current I_(Out). Load 26 may be any electronic device, but isoften a computing device or a communications device, e.g., a computer, acommunications satellite, cellular equipment, or the like.

Power supply 24 is coupled between power source 22 and load 26, and isconfigured to convert D-C energy from the first form supplied by powersource 22 into the second form required by load 26.

The parameters of load 26 may change, often abruptly, and oftensignificantly. For example load 26 may be or include a computer having aprocessor demanding a significant current, as well as auxiliary devices,e.g., a motor and/or a subprocessor, either of which also demandssignificant current, and either or both of which may be instantaneouslyactivated or deactivated to fulfill a given task. Such changes mayresult in transients, i.e., abrupt and significant shifts in outputcurrent I_(Out). Being abrupt, these transients affect output currentI_(Out) dynamically (i.e., during the change) and statically (after thechange). From a dynamic (A-C) perspective, load 26 may be said to havean impedance Z, where the dynamic value of output current I_(Out) at agiven instant is.

$\begin{matrix}{I_{Out} = {\frac{V_{Out}}{Z}.}} & (1)\end{matrix}$From a static (D-C) perspective, impedance Z includes a resistance R,where the static value of output current I_(Out) at a given time is.

$\begin{matrix}{I_{Out} = {\frac{V_{Out}}{R}.}} & (2)\end{matrix}$

Those skilled in the art will appreciate that, since it is often nearlyimpossible to predict the energy demands of load 26 for any given pointin time, power source 22 and power supply 24 are normally designed tomeet a range of output energy demands, from a predetermined minimum to apredetermined maximum, thereby encompassing the requirements of load 26.

In the preferred embodiment, power supply 24 is a sliding-mode switchingpower supply 24 (i.e., a second-order hysteretic switching power supply)configured to receive D-C input voltage V_(In) from power source 22 andto supply D-C output voltage V_(Out) to load 26.

Power supply 24 is configured to divide input voltage V_(In) into Nphases 28, where N is a positive integer. System 20 includes asingle-phase system 20 when N=1 and a multiphase system 20 when N>1.System 20 is assumed to have at least one phase 28. Power supply 24contains N switches 30, where each switch 30 effects one of the N phases28. The N switches 30 are coupled to N inductances 32 in a one-to-onecorrespondence. Each switch 30 alternately connects its particularinductance 32 between power source 22 and a common or ground 34. All Ninductances 32 couple to a capacitance 36. Load 26 couples to the Ninductances 32 and across capacitance 36.

Switches 30 are typically realized as pairs of MOSFETs or other activedevices operating as double-throw switches 30, and make connections ineither throw. For the sake of convention, however, this discussion willassume that a given switch 30 is “on” when it connects its inductance 32to power source 22 and “off” when it connects its inductance 32 toground 34.

FIG. 4 shows a block diagram of a monitor circuit 38 in accordance witha preferred embodiment of the present invention. The followingdiscussion refers to FIGS. 3 and 4.

In the preferred embodiment, monitor circuit 38 is coupled to each ofthe N inductances 32 and to capacitance 36. Monitor circuit 38 monitorsstate variables for power supply 24. As a minimum, monitor circuit 38monitors a capacitive voltage V_(C) (i.e., a voltage across capacitance36). Since capacitance 36 is coupled directly across load 26, capacitivevoltage V_(C) is also output voltage V_(Out).

If, as in the preferred embodiment, monitor circuit 38 monitors nothingmore than output voltage V_(Out), monitor circuit 38 may be implementedas nothing more than a conductor coupling capacitance 36 to a controlcircuit 40. In alternative embodiments, however, monitor circuit 38 mayinclude sensing devices, shown as dotted-line circles in FIG. 4, tomonitor inductive currents I_(L1) through I_(LN) flowing through each ofinductances 32, a capacitive current I_(C) flowing through capacitance36, or output current I_(Out) flowing through load 26.

FIG. 5 shows a block diagram of control circuit 40 configured inaccordance with a preferred embodiment of the present invention. Thefollowing discussion refers to FIGS. 3, 4, and 5.

Control circuit 40 incorporates a first state-variable generator 42, asecond state-variable generator 44, a feedback-signal generator 46, apulse-width-modulation (PWM) generator 48, and a phase selector 50.Optionally, a transient control 52 may be included in some embodiments.Each of these components of control circuit 40 is discussed in detailhereinafter.

Control circuit 40 causes power supply 24 to provide output voltageV_(Out) and output current I_(Out) required by load 26. This isaccomplished by controlling the timing of the outputs 54 of phaseselector 50, which couple to the N switches 30. Control circuit 40therefore controls the switching of the N switches 30 to produce the Nphases 28.

In a preferred embodiment, control circuit 40 receives output voltageV_(Out), which is also capacitive voltage V_(C), from monitor circuit38. Output voltage V_(Out) is routed to first state-variable generator42.

Within first state-variable generator 42, output voltage V_(Out) issubtracted from a reference voltage V_(Ref) by a subtraction circuit 56.Subtraction circuit 56 may be implemented as a simple differenceamplifier, though this is not a requirement of the present invention.

Desirably, reference voltage V_(Ref) is a constant value during normaloperation, and is equal to the desired voltage to be produced by powersupply 24. Reference voltage V_(Ref) need supply only a small amount ofcurrent, and can therefore be easily generated to a desired accuracyusing a wide variety of techniques well known to those skilled in theart.

An output of subtraction circuit 56 is the difference between referencevoltage V_(Ref) and output voltage V_(Out). The output of subtractioncircuit 56 is therefore an analog error voltage V_(E) that represents adifference between output voltage V_(Out) (the actual output voltage)and reference voltage V_(Ref) (the desired output voltage). Thus, duringnormal operation any deviation of output voltage V_(Out) from referencevoltage V_(Ref) represents an error from the desired voltage in theamount of the deviation.

An analog-to-digital (A/D) converter 58 then converts analog errorvoltage V_(E) into a digital error voltage x₁.x ₁ =V _(E) =V _(Out) −V _(Ref).  (3)This digital error voltage x₁ is a first (voltage) state variable x₁ ofpower supply 24.

It will be appreciated by those skilled in the art that in alternativeembodiments other signals from monitor circuit 38 may be routed to firststate-variable generator 42 for processing in other manners notdiscussed herein. These embodiments are represented by a dotted line 60in FIG. 5. Such an alternative signal may correspond to output currentI_(Out) flowing through load 26, correspond to capacitive current I_(C)flowing through capacitance 36, or independently correspond to inductivecurrents I_(L1) through I_(LN) flowing in one or more of inductances 32.The use of these or other alternative signals does not depart from thespirit of the present invention. In the preferred embodiment, however,collective and individual inductive currents are estimated from firststate variable x₁, as discussed in more detail hereinafter.

In the preferred embodiment, A/D converter 58 is a small (4-bit), fast(50 MHz) converter. This is a simple and inexpensive approach. Thoseskilled in the art will appreciate that other A/D converters may be usedwithout departing from the spirit of the present invention.

Subsequent to A/D converter 58, the circuits and functions of controlcircuit 40 may be implemented using digital hardware logic and/ormicroprocessor circuits, the design and logic of which can vary widelyfrom application-to-application but which can be readily adapted bythose skilled in the digital electronic arts.

In the preferred embodiment, first state variable x₁, derived fromoutput voltage V_(Out), is the only state variable that needs to bederived directly from a physically monitored parameter of power supply24. Any other state variable is derived by calculation from first statevariable x₁.

First state variable x₁ is output from first state-variable generator 42and routed to second-state-variable generator 44 and feedback-signalgenerator 46. Within second state-variable generator 44, first statevariable x₁ passes to an inductive-current generator 62.Inductive-current generator 62 calculates a second state variable x₂ asan error rate x₂ (i.e., a rate of error voltage x₁) by extracting thederivative of first state variable x₁ over time.

$\begin{matrix}{x_{2} = {\frac{\mathbb{d}x_{1}}{\mathbb{d}t} = {\frac{\mathbb{d}V_{Out}}{\mathbb{d}t} = {\frac{I_{C}}{C}.}}}} & (4)\end{matrix}$This is equivalent to taking the derivative of output voltage V_(Out)over time, and is substantially equal to capacitive current I_(C)divided by a value C of capacitance 36.

Inductive-current generator 62 also derives signals representing each ofinductive currents I_(L1) through I_(LN), which signals are effectivelysynthesized inductive currents. Inductive-current generator 62 derives afirst synthesized inductive current representing a first inductivecurrent I_(L1) through a first inductance 32′ during a first phase 28′,a second synthesized inductive current representing a second inductivecurrent I_(L2) through a second inductance 32″ during a second phase28″, and so on through an N^(th) synthesized inductive currentrepresenting an N^(th) inductive current I_(LN) through an N^(th)inductance 32 ^(N) during an N^(th) phase 28 ^(N). Each synthesizedinductive current represents one of the N inductive currents I_(L1)through I_(LN). Therefore, this discussion refers to either synthesizedor physical inductive currents as inductive currents I_(L1) throughI_(LN).

Inductive-current generator 62 may receive a signal from phase selector50 (discussed hereinafter) to identify the currently active phase 28.

Those skilled in the art will appreciate that it is not necessary forinductive-current generator 62 to generate signals that equal inductivecurrents I_(L1) through I_(LN) as long as the signals generated bear thesame relationships to each other as do inductive currents I_(L1) throughI_(LN).

In an alternative embodiment, inductive current generator 62 maydirectly receive one or more of inductive currents I_(L1) through I_(LN)from A/D converter 58, or may associate output current I_(Out) with thecurrently active phase 28 to parse out inductive currents I_(L1) throughI_(LN) for each phase 28.

Feedback-signal generator 46 generates a single feedback signal σ.Moreover, feedback-signal generator 46 produces single feedback signal σfrom no more than two state variables of power supply 24, error voltagex₁ and error rate x₂. In the preferred embodiment, single feedbacksignal σ controls all N phases 28. Accordingly, single feedback signal σis a comprehensive feedback signal σ because it influences all N phases28 of multiphase power supply 24.

FIG. 6 shows a phase plot depicting a sliding surface σ and a variablewindow Δσ therefor in accordance with a preferred embodiment of thepresent invention. The following discussion refers to FIGS. 3, 4, 5, and6.

In the preferred embodiment, feedback-signal generator 46 within controlcircuit 40 is a sliding-surface generator 78 that generatescomprehensive feedback signal a in the form of a single sliding surfaceσ. By generating comprehensive feedback signal σ as single slidingsurface σ, the number of state variables required by feedback-signalgenerator 46 may be kept to a minimum (i.e., two. one monitored, onecalculated) regardless of the number N of phases 28. This small numberof state variables is desirable because it leads away from the rapidlyexpanding complexity encountered in conventional multiphase powersupplies that monitor N+1 state variables to control N phases 28. Theuse of a small number of state variables therefore improves reliabilityand decreases expense over traditional methodologies.

Hereinafter in this discussion, feedback-signal generator 46 andfeedback signal σ are referred to as sliding-surface generator 78 andsliding surface σ, respectively. Sliding-surface generator 78 isdiscussed in more detail hereinafter.

FIG. 6 depicts the state variables as error voltage x₁ on the horizontalaxis and error rate x₂ (i.e., synthesized current) on the vertical axis.As such, FIG. 6 depicts the operation of power supply 24 as defined byits state variables x₁ and x₂. Two spirals are depicted on thehorizontal axis, with one spiral centered about a positive value where.x ₁ =V _(Out) −V _(Ref) ,x ₂=0,  (5)and another spiral centered around a negative value where.x ₁ =−V _(Ref) ,x ₂=0.  (6)The positive-value spiral depicts an exemplary track state variables x₁and x₂ might follow if switches 30 were continuously on, and thenegative-value spiral depicts an exemplary track state variables x₁ andx₂ might follow if switches 30 were continuously off. Of course,switches 30 are neither continuously on nor continuously off, but areswitched on and off with the goal of maintaining a value of the statevariables where:x₁=0, x₂=0.  (7)It is a task of sliding-surface generator 78 to identify when suchswitching should take place, although sliding surface σ generated bysliding-surface generator 78 may be adjusted as discussed herein by acurrent-balance control 80 and a variable-window generator 82.

Sliding-mode controls are known to those skilled in the art.Sliding-surface generator 78 is a sliding-mode control that has beenadapted for use with power supply 24. In the preferred embodiment,sliding-surface generator 78 generates single sliding surface σ as afunction.σ=α·x ₁ +x ₂,  (8)where α is a constant. First state variable (error voltage) x₁ is amonitored voltage state variable, and second state variable (error rate)x₂ is a derived (synthesized) current state variable.

The goals used in establishing this relationship are known to thoseskilled in the art of sliding-mode controls. In general, error voltagex₁, error rate x₂ (the rate of change of error voltage x₁ over time),and even the acceleration of error voltage x₁ in time may all be takeninto account in defining sliding surface σ. Of course, those skilled inthe art will appreciate that sliding surface σ is an idealized result.In practice, the state of power supply 24 will seldom be precisely onsliding surface σ. Rather, switches 30 are controlled so that futureoperation of power supply 24 will, (except for the operations ofcurrent-balance control 80 and variable-window generator 82), bedirected toward sliding surface σ and the origin of the phase plot shownin FIG. 6. By thus controlling the activation and deactivation ofswitches 30, the operation of power supply 24, as demonstrated by itsstate variables x₁ and x₂, will tend to “slide” along sliding surface σ.

In the preferred embodiment, constant α has a range.

$\begin{matrix}{{0 \leq \alpha \leq \frac{1}{\tau}},} & (9)\end{matrix}$where τ is a time constant.τ==R·C,  (10)where C is the value of capacitance 36 and R is the resistance of load26 (i.e., the resistive component of load impedance Z).

As discussed hereinbefore in conjunction with equations (3) and (4),first state variable (error voltage) x₁ represents a difference betweenoutput voltage V_(Out) and reference voltage V_(Ref), and second statevariable (error rate) x₂ represents a rate of change of first statevariable x₁ and is the derivative thereof.

By physically measuring only first state variable x₁ and simulatingsecond state variable x₂ from first state variable x₁, only a singlesliding surface (comprehensive feedback signal) σ is required to controlany number N of phases 28. Moreover, by refraining from physicallymeasuring a current state variable, no lossy current-measuring devicesare required. This further improves reliability and decreases cost inaddition to improving efficiency.

Sliding-surface generator 78 generates sliding surface σ, a signal thatsummarizes and describes the operating state of power supply 24. Thoseskilled in the art will appreciate that sliding surface σmay also becalled a sliding or switching line, curve, plane, or hyperplane in othersliding-mode control applications.

PWM generator 48 is coupled to feedback-signal generator 46. PWMgenerator 48 is configured to translate single sliding surface σ into aPWM signal 84 consisting of a stream of switching pulses 86. Phaseselector 50 (discussed hereinafter) routes different switching pluses 86within PWM signal 84 to different switches 30.

The following discussion refers to FIGS. 3 and 5.

Within PWM generator 48, optional current-balance control 80 adjustssliding surface σ and alters it into an adjusted sliding surface σ′.Current-balance control 80 receives signals from inductive-currentgenerator 62 corresponding to inductive currents I_(L1) through I_(LN)flowing in each of inductances 32.

Current-balance control 80 receives inputs that correspond to the Ninductive currents I_(L1) through I_(LN) for each phase 28. For anM^(th) one of the N phases 28, where M is an integer in the range 1≦M≦N,the M^(th) inductive current I_(LM) is a current through an M^(th) oneof the N inductances 32 coupled to an M^(th) one of the N switches 30effecting the M^(th) phase 28. Since the M^(th) inductive current I_(LM)is germane only to the M^(th) phase 28, the M^(th) inductive currentI_(LM) is a phase current for that M^(th) phase 28.

It is desirable that all of the N inductive currents I_(L1) throughI_(LN) be substantially equal so that power supply 24 can supply themaximum current within the capacity of a given set of switches 30,thereby maximizing overall efficiency and reliability. Current-balancecontrol 80 computes a summary statistic I_(X) (not shown) as a referencecurrent for the N inductive currents I_(L1) through I_(LN). In thepreferred embodiment, summary statistic I_(X) is desirably an arithmeticmean of the N inductive currents I_(L1) through I_(LN).

$\begin{matrix}{I_{X} = {\frac{I_{L\; 1} + I_{L\; 2} + \ldots + I_{LN}}{N}.}} & (11)\end{matrix}$Those skilled in the art will appreciate that this is not a requirementof the present invention, and that summary statistic I_(X) may be otherthan the arithmetic mean without departing from the spirit of thepresent invention.

For each phase 28, current-balance control 80 then computes an errorcurrent I_(E) (not shown) as a difference between summary statisticI_(X) and an inductive current I_(L) for that phase 28.I _(E) =I _(Ref) −I _(L).  (12)For each M^(th) phase 28, current-balance control 80 then alters slidingsurface σ into adjusted sliding surface σ′ so that inductive currentI_(L) for that phase 28 is substantially equal to summary statisticI_(X).

In particular, current-balance control 80 adds an offset proportional toerror current I_(E) (not shown) to sliding surface σ when a given phase28 has been providing inductive current I_(L) not equal to summarystatistic I_(X). The provided offset will make adjusted sliding surfaceσ′ slightly different from sliding surface σ, and the operation of powersupply 24 for that phase 28 will slide along adjusted sliding surfaceσ′. In this manner, all phase currents I_(L1) through I_(LN) arerendered substantially equal.

Current-balance control 80 is a desirable but optional component incontrol circuit 40. This discussion assumes the presence ofcurrent-balance control 80. If current-balance control 80 is omitted,then sliding surface σ is not adjusted to become adjusted slidingsurface σ′, and any mention of sliding surface σ hereinafter alsoapplies to adjusted sliding surface σ′.

The following discussion refers to FIGS. 3, 5, and 6.

Adjusted sliding surface σ′ is routed to a translation circuit 102configured to convert adjusted sliding surface σ′ into PWM signal 84,wherein PWM signal 84 consists of a stream of switching pulses 86 atsubstantially a switching frequency f_(S).

Variable-window generator 82 is configured to compare sliding surface σto two offset values. In particular, variable-window generator 82bifurcates sliding surface σ. Whenever operation greater than a highthreshold 106 of sliding surface σ is detected, variable-windowgenerator 82 activates, causing a switch 30 to switch on. This effectsone of the N phases 28. Whenever operation less than a low threshold 108of sliding surface σ is detected, variable-window generator 82deactivates, causing the currently active switch 30 to switch off. Thoseskilled in the art will appreciate that other methodologies for theoperation of variable-window generator 82 may be used without departingfrom the spirit of the present invention.

FIG. 6 also depicts an exemplary oscillating sliding signal 110 betweenthe limits of high threshold 106 and low threshold 108. Sliding signal110 depicts the operation of power supply 24, as defined by its statevariables x₁ and x₂, as it slides along sliding surface σ. Oscillationresults from switching switches 30 at switching frequency f_(S). Thus,the oscillation frequency of sliding signal 110 tracks switchingfrequency f_(S).

Switching frequency f_(S) produces a ripple frequency f_(R) acrosscapacitance 36, and therefore across load 26. It is desirable thatripple frequency f_(R) be substantially fixed or constant so that anyinterference and/or harmonic effects produced thereby may more easily besuppressed within the electronic device serving as load 26. A constantpredetermined frequency f_(P) (not shown) serves as an ideal or targetripple frequency f_(R). That is, power supply 24 in general, andconstant-frequency control 104 in particular, maintain switchingfrequency f_(S), and therefore ripple frequency f_(R), substantiallyequal to constant predetermined frequency f_(P).

Within translation circuit 102, a reference generator 112 generates afixed reference frequency f_(X), and a frequency comparator 114 comparesswitching frequency f_(S) against reference frequency f_(X) to produce afrequency error E_(f). In the preferred embodiment, reference generator112 provides fixed reference frequency f_(X) as a clock signalexhibiting a frequency much greater than the expected switchingfrequency f_(S). Frequency comparator 114 counts reference frequencyf_(X) for a predetermined number of cycles of switching frequency f_(S).The resultant count then becomes frequency error E_(f). As shown in FIG.6, an increase in the count indicates a decrease in switching frequencyf_(S), shown as a frequency-decreasing direction (arrow) 116 and adecrease in the count indicates an increase in switching frequencyf_(S), shown as a frequency-increasing direction (arrow) 118. Thoseskilled in the art will appreciate that other methodologies may be usedto determine frequency error E_(f). The use of another methodology doesnot depart from the spirit of the present invention.

Frequency error E_(f) is applied to variable-window generator 82.Variable-window generator 82 forms and/or adjusts variable window Δσ inresponse to frequency error E_(f). In the preferred embodiment,frequency error E_(f) is scaled by an appropriate multiplier and addedto a constant corresponding to the desired predetermined constantfrequency to produce an offset. The offset thus produced increases whenan increase 118 in switching frequency f_(S) is detected, andvice-versa. Those skilled in the art will appreciate that othermethodologies for adjusting variable window Δσ may be used withoutdeparting from the spirit of the present invention.

Constant-frequency control 104 maintains switching frequency f_(S)substantially equal to predetermined constant frequency under staticconditions. Unfortunately, conditions are dynamic, not static. Load 26exhibits changes, sometimes abruptly, and sometimes significantly.Switching frequency f_(S) is dependent not only upon variable window Δσ,but also upon the demands of load 26. Specifically, if impedance Z(i.e., resistance R) of load 26 changes, output current I_(Out) changesand switching frequency f_(S) is urged to change.

When load 26 exhibits a change, output current I_(Out) instantly changesas well. The change in output current I_(Out) produces a first marginalchange in switching frequency f_(S). The first marginal change may be infrequency-increasing direction 116 for an increase in output currentI_(Out) or a frequency-decreasing direction 118 for a decrease in outputcurrent I_(Out). The degree or amount of change is not relevant for thepurposes of this discussion. Frequency comparator 114 compares themarginally changed switching frequency f_(S) against reference frequencyf_(X) and produces frequency error E_(f) reflecting the change inswitching frequency f_(S). Constant-frequency control 104 then adjustsvariable window Δσ accordingly by means of an offset to sliding surfaceσ. This adjustment produces a second marginal change in switchingfrequency f_(S) in opposition to the first marginal change. In otherwords, if the first marginal change was in frequency-decreasingdirection 118, then the second marginal change will be infrequency-increasing direction 116, and vice-versa. This second marginalchange causes switching frequency f_(S) to become more closely equal tothe desired predetermined constant frequency.

Those skilled in the art will appreciate that the second marginal changeneed not completely offset the first marginal change. Indeed, tocompensate for the first marginal change in a single step is contrary toaccepted negative feedback practice and may lead to instability.Successive iterations of the process are normally used, where eachiteration renders switching frequency f_(S) more closely equal to thedesired predetermined constant frequency. This practice will rapidlybring and maintain switching frequency f_(S) into substantial equalitywith the desired predetermined constant frequency.

Constant-frequency control 104 will maintain switching frequency f_(S)substantially constant. Those skilled in the art will appreciate that itneed not maintain perfect constancy but desirably maintains substantialconstancy within a reasonable tolerance about predetermined constantfrequency.

Transients may occur in response to sudden and/or significant changes inload 26. Under certain conditions, these transients may drive powersupply 24 into a lockup condition for an undesirably long duration.Eventually, the lockup condition will clear, even when inductances 32,capacitance 36, and/or load 26 are vastly mismatched and/or vary widelyfrom design ideals.

During such a transient, feedback-signal generator 46 may output slidingsurface σ that might, when further processed by translation circuit 102,instruct the switch 30 effecting the currently active phase 28 to remainin its on condition. This is may lead to a failure of that switch 30,and is therefore undesirable.

PWM signal 84 is output from PWM generator 48, and input to an optionaltransient control 52. Within transient control 52, PWM signal 84 ismodified to become a modified PWM signal 124. Modified PWM signal 124 isconfigured to regulate a duration of a lockup condition of any givenswitch 30 due to a transient from load 26. By regulating the duration ofa lockup condition, transient control 52 inhibits potential damage tothat switch 30, and to power supply 24.

In the event of a lockup condition, transient control 52 causes thecurrently active phase 28 to terminate and the next phase 28 to commenceafter a predetermined duration. In other words, switches 30 are switchedfrom the currently active phase 28 to the next phase 28. Lockupconditions are thereby tolerated.

Transient control 52 is a desirable but optional component in controlcircuit 40. This discussion assumes the presence of transient control52, and that PWM signal 84 becomes modified PWM signal 124. If transientcontrol 52 is omitted, then PWM signal 84 is not modified, i.e.,modified PWM signal 124 is PWM signal 84.

FIG. 7 shows a chart depicting the distribution and switching ofswitching pulses 86 to each of switches 30 for the N phases 28 inaccordance with a preferred embodiment of the present invention. For thesake of simplicity, FIG. 7 assumes N phases 28 where N=3 (i.e., threephases 28). Those skilled in the art will appreciate that this isexemplary only, applies only to multiphase systems 20, and that N may beany desired integer greater than one. The following discussion refers toFIGS. 3, 4, 5, and 7.

Within phase selector 50, PWM signal 84 is routed to a phase counter136. Phase counter 136 identifies which of the N phases 28 is to beactive at any given point in time. Desirably, phase counter 136 “counts”once for each switching pulse 86 in PWM signal 84, cycling as required.For example, in a three-phase application (i.e., where N=3, as in FIG.7), phase counter 136 may count from zero to two, then on the receipt ofthe next switching pulse from PWM signal 84 be reset back to zero.

In the preferred embodiment, phase counter 136 is realized as a ringcounter. Those skilled in the art will appreciate that this is not arequirement of the present invention, and that other embodiments ofphase counter 136 may be used without departing from the spirit of thepresent invention.

Output from phase counter 136 are provided to a switching circuit 138,as well as to inductive-current generator 62 and to current-balancecontrol 80 as discussed hereinbefore. This allows inductive-currentgenerator 62 and current-balance control 80 to coordinate theiractivities with the currently active phase 28.

Switching circuit 138 sequentially distributes switching pulses 86 fromPWM signal 84 to the control inputs of the N switches 30. Accordingly,the output from phase counter 136 identifies the target switch 30 to beswitched on or off, and PWM signal 84 provides the timing for theswitch-on and switch-off events. Signals 54 from switching circuit 138couple to control inputs of each of switches 30 in each of phases 28.That is, a first signal 54′ is coupled to the control inputs of a firstswitch 30′ to effect first phase 28′, a second signal 54″ is coupled tothe control inputs of a second switch 30″ to effect second phase 28″,and so forth until an N^(th) signal 54 ^(N) is coupled to the controlinputs of an N^(th) switch 30 ^(N) to effect the N^(th) phase 28 ^(N).

Collectively, switches 30 then switch at switching frequency f_(S). Foran M^(th) one of the N phases 28, switching circuit 138 switches from anM^(th) to an (M+1)^(th) switch 30 under the control of phase counter136.

In a multiphase power supply 24 (i.e., when N>1), as in the preferredembodiment, switching circuit 138 may be implemented using a relativelysimple decoding circuit (not shown) well known to those skilled in theart. It will be appreciated, however, that alternative implementationsof switching circuit 138 may be used without departing from the spiritof the present invention.

Those skilled in the art will appreciate that for a single-phase powersupply 24 (i.e., when N=1), transient control 52 and phase counter 136may be omitted. In this situation, switching circuit 138 may beimplemented using conductors or buffering circuits which route PWMsignal 84 to the control inputs of switch 30 for that single phase 28.

In summary, the present invention teaches an apparatus and method forfixed-frequency control in a switching power supply 24. Power supply 24has a control circuit 40, suitable for either single-phase or multiphasesystems 20, that is simple, reliable, and inexpensive, and requires onlytwo state variables x₁ and x₂, regardless of the number N of phases 28.Power supply 24 is substantially independent of component tolerance, istolerant of lockup conditions of switches 30, and has a switchingfrequency f_(S) relatively immune to variations in load 26.

Although the preferred embodiments of the invention have beenillustrated and described in detail, it will be readily apparent tothose skilled in the art that various modifications may be made thereinwithout departing from the spirit of the invention or from the scope ofthe appended claims.

1. A fixed-frequency switching power supply comprising: N switchescoupled to a D-C power source and configured to receive an input energytherefrom, wherein N is a positive integer, and wherein said N switchesare configured to divide said input energy into N phases; N inductances,wherein each of said N inductances is coupled to one of said N switches;a capacitance coupled to each of said N inductances and coupled to adynamic load, wherein said capacitance and said N inductances aretogether configured to provide an output energy to said dynamic load; amonitor circuit coupled to said capacitance and configured to monitorsaid output energy; and a control circuit coupled to said monitorcircuit, configured to derive a first state variable of said powersupply from said monitor circuit, to synthesize a second state variableof said power supply from said first state variable, and to switch saidN switches at a substantially constant switching frequency in responseto said first and second state variables.
 2. A power supply as claimedin claim 1 wherein. said input energy is D-C energy in a first formcomprising an input current at an input voltage; said output energy isD-C energy in a second form comprising an output current at an outputvoltage; and said input voltage does not equal said output voltage.
 3. Afixed-frequency switching power supply comprising: N switches coupled toa D-C power source and configured to receive an input energy therefrom,wherein N is a positive integer, and wherein said N switches areconfigured to divide said input energy into N phases; N inductances,wherein each of said N inductances is coupled to one of said N switches;a capacitance coupled to each of said N inductances and coupled to adynamic load, wherein said capacitance and said N inductances aretogether configured to provide an output energy to said dynamic load; amonitor circuit coupled to said capacitance and configured to monitorsaid output energy; and a control circuit coupled to said monitorcircuit and configured to switch said N switches at a substantiallyconstant switching frequency in response to said monitor circuit,wherein said control circuit comprises: a first state-variable generatorcoupled to said monitor circuit and configured to generate a first statevariable; a second state variable generator coupled to said firststate-variable generator and configured to synthesize a second statevariable; a sliding-surface generator coupled to said monitor circuitand configured to generate a single sliding surface for said N phases inresponse to said first and second state variables; apulse-width-modulation (PWM) generator coupled to said sliding-surfacegenerator and configured to translate said single sliding surface into astream of switching pulses at said substantially constant switchingfrequency; and a phase selector coupled to said N switches, coupled tosaid PWM generator, and configured to switch said N switches at saidsubstantially constant switching frequency in response to said stream ofswitching pulses so that each of said N switches effects one of said Nphases.
 4. A power supply as claimed in claim 3 wherein said slidingsurface has a value αx₁+x₂, wherein: α is a constant having a range,0≦α≦(τ)−1, where τ is a time constant having a value RC, where R is avalue of a resistive component of said load; and C is a value of saidcapacitance; x₁ is a first two state variable and has a valueV_(Ref)−V_(Out), where V_(Ref) is a value of a reference voltage, andV_(Out) is a value of an output voltage of said power supply; and x₂ isa second state variable and is a derivative of said first state variableover time.
 5. A power supply as claimed in claim 3 wherein: said firststate variable is a difference between a voltage of said output energyand a fixed reference voltage equal to an idealized output voltage ofsaid power supply; and said second state variable is a derivative ofsaid first state variable over time.
 6. A power supply as claimed inclaim 3 wherein: said second state-variable generator is an inductivecurrent generator configured to synthesize N synthesized inductivecurrents from said first state variable; and said PWM generatorcomprises a current-balance control coupled to said sliding-surfacegenerator, coupled to said inductive current generator, and configuredto adjust said comprehensive feedback signal so that N inductivecurrents are rendered substantially equal in response to said Nsynthesized inductive currents, wherein each of said N inductivecurrents is a current through one of said N inductances.
 7. A powersupply as claimed in claim 3 wherein: said PWM generator is configuredto form a variable window for said sliding surface, and said phaseselector is configured to switch said N switches at said switchingfrequency as determined by said variable window.
 8. A power supply asclaimed in claim 7 wherein said switching frequency is determined bysaid variable window and said load.
 9. A power supply as claimed inclaim 8 wherein: a change in said dynamic load produces a first marginalchange in said switching frequency; and said PWM generator produces asecond marginal change in said switching frequency in opposition to saidfirst marginal change.
 10. A power supply as claimed in claim 3 wherein:said phase selector collectively switches said N switches at saidsubstantially constant switching frequency; and said phase selectorindividually switches each of said N switches at substantially N⁻¹ timessaid substantially constant switching frequency.
 11. A method ofoperating a fixed-frequency switching power supply, said methodcomprising: receiving an input energy from a D-C power source; providingan output energy to a dynamic load; monitoring said output energy;deriving a first state variable of said power supply in response to saidmonitoring activity; synthesizing a second state variable of said powersupply in response to said deriving activity; producing a stream ofswitching pulses at a substantially constant frequency in response tosaid deriving and synthesizing activities; switching N switches inresponse to said stream of switching pulses, where N is a positiveinteger; and effecting N phases of said input D-C energy with said Nswitches.
 12. A method as claimed in claim 11 wherein said derivingactivity comprises: generating a fixed reference voltage; andsubtracting an output voltage of output energy from said fixed referencevoltage.
 13. A method as claimed in claim 11 wherein said synthesizingactivity comprises extracting a derivative of said first state variableover time.
 14. A method of operating a fixed-frequency switching powersupply, said method comprising: receiving an input energy from a D-Cpower source; providing an output energy to a dynamic load; monitoringsaid output energy; producing a stream of switching pulses at asubstantially constant frequency in response to said monitoringactivity, wherein said producing activity comprises; generating asliding surface from no more than two state variables of said powersupply; forming a variable window for said sliding surface; andtranslating said sliding surface into a stream of switching pulses;switching N switches in response to said stream of switching pulses,where N is a positive integer; and effecting N phases of said input D-Cenergy with said N switches.
 15. A method as claimed in claim 14 whereinsaid generating activity generates said sliding surface for said Nphases, said sliding surface having a value αx₁+x₂, where: α is aconstant not less than zero; x₁ is a first state variable of said powersupply; and x₂ is a second state variable of said power supply.
 16. Amethod as claimed in claim 14 additionally comprising: generatingsignals corresponding to inductive currents through each of Ninductances in response to said monitoring activity, wherein saidsignals are N synthesized inductive currents; and adjusting said slidingsurface for each of said N phases so that said N synthesized inductivecurrents are rendered more closely equal.
 17. A method as claimed inclaim 11 additionally comprising: producing a first marginal change insaid switching frequency in response to a change in said load; andproducing a second marginal change in said switching frequency inopposition to said first marginal change.
 18. A system comprising: a D-Cpower source; a dynamic load; and a fixed-frequency sliding-modeswitching power supply having N phases, where N is a positive integer,and configured to receive an input energy from said D-C power source, toconvert said input energy into an output energy, and to provide saidoutput energy to said dynamic load, said power supply comprising: Nswitches coupled to said D-C power source, wherein each of said Nswitches effects one of said N phases; N inductances, wherein each ofsaid N inductances is coupled to one of said N switches; a capacitancecoupled to said N inductances and said dynamic load; a monitor circuitcoupled to said capacitance and configured to monitor said outputenergy; and a sliding mode control circuit coupled to said monitorcircuit, having a sliding surface generator configured to generate asingle sliding surface for said N phases as said feedback signal, andconfigured to switch said N switches at a substantially constantswitching frequency in response to said monitor circuit.
 19. A system asclaimed in claim 18 wherein said sliding-mode control circuit comprises:a constant-frequency control coupled to said sliding-surface generator,configured to translate said sliding surface into a stream of switchingpulses at a substantially constant switching frequency, and comprising:a reference generator configured to generate a reference frequency; afrequency comparator configured to compare said switching frequency tosaid reference frequency to produce a frequency error; and avariable-window generator configured to form a variable window for saidsliding surface in response to said error frequency; and a phaseselector coupled to said N switches, coupled to said constant-frequencycontrol, and configured to sequentially switch said N switches at saidsubstantially constant switching frequency in response to said stream ofswitching pulses, wherein: said phase selector switches said N switchesin response to said sliding surface, said variable window, and animpedance of said dynamic load; said dynamic load produces a firstmarginal change in said switching frequency in response to a change insaid impedance; and said constant-frequency control adjusts saidvariable window to produce a second marginal change in said switchingfrequency in opposition to said first marginal change.